The simplest low-frequency amplifiers using transistors. Transistor amplifier: types, circuits, simple and complex Practical circuit of a four-stage transistor amplifier

ASSIGNMENT FOR A COURSE PROJECT

Develop a two-stage amplifier circuit with direct coupling.

The initial data for design are given in Table 1.

Table 1. Initial data

INTRODUCTION

ANALYTICAL PART

2. Selection of transistor by cutoff frequency, maximum collector-emitter voltage and maximum collector current

3. Calculation of the DC operating mode of the transistor and selection of passive circuit elements: resistors, capacitors, inductances

4. Calculation of the circuit according to alternating current, consisting of determining the gain, input and output resistance of the stage

5. Calculation of nominal values ​​of passive and frequency-setting circuit elements

6. Replacement of calculated values ​​of passive elements with values ​​from the E24 series

7. Test calculation of the operating mode of the electronic circuit

8. Simulation of circuit operation in the MicroCap 8 environment

CONCLUSION

LIST OF REFERENCES USED

INTRODUCTION

The purpose of this course project is to study the methodology for setting a problem when designing electrical circuit diagrams on semiconductor devices, drawing up a technical specification for the designed device, gaining skills in the phased integrated circuit design of electrical components, gaining experience in using modern information technologies and simulation systems.

In this course project, a two-stage amplifier circuit with direct coupling is developed.

ANALYTICAL PART

1. Selecting an electronic device circuit depending on the specified parameters

Rice. 1. Circuit of a two-stage amplifier with direct coupling.

The selection of the circuit of the electronic device is carried out in accordance with the received task and the characteristics of the device.

Since it is necessary to provide a high gain and there is no need for a very high input resistance, we will choose the OE-OE circuit.

According to the assignment option, a two-stage amplifier circuit with direct coupling according to the OE-OE circuit was selected (Fig. 1.)

2. Selection of the transistor of the second stage according to the cutoff frequency, maximum collector-emitter voltage and maximum collector current

The main criterion for choosing the type of transistor for the amplifier stage is the permissible voltage between the collector and emitter U CE, which is determined from the condition

The maximum collector current of the transistor must exceed the operating current of the cascade

The cutoff frequency of the transistor must exceed the maximum frequency of the operating range DF

Based on the results obtained, we select the transistor KT312V (VF240). For the selected bipolar transistor, we write out the reference data and enter them in Table 2.

Table 2.

3. Calculation of the operating mode of the second stage transistor for direct current and selection of passive circuit elements: resistors, capacitors, inductors

We begin the calculation by selecting the quiescent current of the bipolar transistor I K 0 . Since the cascade operates in mode A, the collector current is selected from the ratio:

For the proposed amplifier circuit, R H is not specified, so we select the collector current equal to 45 mA.

Figure 2 shows the family of transistor output characteristics.

Let us determine the position of the operating point on the output characteristic of the selected transistor, taking into account that it operates in mode A.

Rice. 2. Family of output characteristics of the KT312V transistor

Since the transistor operates in mode A, then U KO = E P / 2 = 12 / 2 = 6 V.

Let's take U KO = 6 V.

The quiescent current of the transistor base is determined from the relationship:

Using two points (I KO, U KO) = (0.045 A, 6 V) and (0, E P) = (0. 12 V) on the family of output characteristics, we construct a load straight line.

We select the operating point at a collector current of 22.5 mA, a collector-emitter voltage of 9 V.

The voltage divider on resistors R K1 R VT 1 R E1 must provide the calculated value of the base current. For this, the condition must be met

then the nominal values ​​of R K 1 and R VT 1 R E1 can be determined from the condition

where U B is selected from the condition U B = U BE + U E – for low-power silicon transistors U BE = 0.5...0.8 V.

For the amplifier stage, U E is usually chosen within the range of (0.1...0.3) E P.

Let's take R K 1 equal to 2861 Ohms. Then U B = 1.7 V.

The current passing through the resistor R E2 is determined by the sum of the collector and base currents

then the nominal value of R E2 can be determined by the formula

The total resistance of the cascade through which the collector current flows is equal to

Calculation of the second stage for alternating current, consisting of determining the gain, input and output resistance of the cascade.

voltage gain

input and output resistance

Output stages based on "twos"

As a signal source we will use an alternating current generator with a tunable output resistance (from 100 Ohms to 10.1 kOhms) in steps of 2 kOhms (Fig. 3). Thus, when testing the VC at the maximum output resistance of the generator (10.1 kOhm), we will to some extent bring the operating mode of the tested VC closer to a circuit with an open feedback loop, and in another (100 Ohm) - to a circuit with a closed feedback loop.

The main types of composite bipolar transistors (BTs) are shown in Fig. 4. Most often in VC, a composite Darlington transistor is used (Fig. 4a) based on two transistors of the same conductivity (Darlington “double”), less often - a composite Szyklai transistor (Fig. 4b) of two transistors of different conductivity with a current negative OS, and even less often - a composite Bryston transistor (Bryston, Fig. 4 c).
The "diamond" transistor, a type of Sziklai compound transistor, is shown in Fig. 4 g. Unlike the Szyklai transistor, in this transistor, thanks to the “current mirror”, the collector current of both transistors VT 2 and VT 3 is almost the same. Sometimes the Shiklai transistor is used with a transmission coefficient greater than 1 (Fig. 4 d). In this case, K P =1+ R 2/ R 1. Similar circuits can be obtained using field-effect transistors (FETs).

1.1. Output stages based on "twos". "Deuka" is a push-pull output stage with transistors connected according to a Darlington, Szyklai circuit or a combination of them (quasi-complementary stage, Bryston, etc.). A typical push-pull output stage based on a Darlington deuce is shown in Fig. 5. If emitter resistors R3, R4 (Fig. 10) of input transistors VT 1, VT 2 are connected to opposite power buses, then these transistors will operate without current cut-off, i.e. in class A mode.

Let's see what pairing the output transistors will give for the two "Darlingt she" (Fig. 13).

In Fig. Figure 15 shows a VK circuit used in one of the professional and onal amplifiers.

The Siklai scheme is less popular in VK (Fig. 18). At the early stages of the development of circuit design for transistor UMZCHs, quasi-complementary output stages were popular, when the upper arm was performed according to the Darlington circuit, and the lower one according to the Sziklai circuit. However, in the original version, the input impedance of the VC arms is asymmetrical, which leads to additional distortion. A modified version of such a VC with a Baxandall diode, which uses the base-emitter junction of the VT 3 transistor, is shown in Fig. 20.

In addition to the considered “twos,” there is a modification of the Bryston VC, in which the input transistors control transistors of one conductivity with the emitter current, and the collector current controls transistors of a different conductivity (Fig. 22). A similar cascade can be implemented on field-effect transistors, for example, Lateral MOSFET (Fig. 24).

The hybrid output stage according to the Sziklai circuit with field-effect transistors as outputs is shown in Fig. 28. Let's consider the circuit of a parallel amplifier using field-effect transistors (Fig. 30).

As effective way To increase and stabilize the input resistance of the “two,” it is proposed to use a buffer at its input, for example, an emitter follower with a current generator in the emitter circuit (Fig. 32).

Of the “twos” considered, the worst in terms of phase deviation and bandwidth was the Szyklai VK.

Let's see what using a buffer can do for such a cascade. If instead of one buffer you use two on transistors of different conductivities connected in parallel (Fig. 35), then you can expect further improvement in parameters and an increase in input resistance. Of all the considered two-stage circuits, the Szyklai circuit with field-effect transistors showed itself to be the best in terms of nonlinear distortions. Let's see what installing a parallel buffer at its input will do (Fig. 37).

The parameters of the studied output stages are summarized in Table. 1 .
Analysis of the table allows us to draw the following conclusions:
- any VC from the “twos” on the BT as a UN load is poorly suited for working in a high-fidelity UMZCH;
- the characteristics of a VC with a DC at the output depend little on the resistance of the signal source;
- VK Siklai with a DC output and a parallel buffer at the input (Fig. 37) has the highest characteristics (minimum distortion, maximum bandwidth, zero phase deviation in the audio range).

Output stages based on "triples"

In high-quality UMZCHs, three-stage structures are more often used: Darlington triplets, Shiklai with Darlington output transistors, Shiklai with Bryston output transistors and other combinations. One of the most popular output stages at present is a VC based on a composite Darlington transistor of three transistors (Fig. 39).

In Fig. Figure 41 shows a VC with cascade branching: the input repeaters simultaneously operate on two stages, which, in turn, also operate on two stages each, and the third stage is connected to the common output. As a result, quad transistors operate at the output of such a VC.

The VC circuit, in which composite Darlington transistors are used as output transistors, is shown in Fig. 43. The parameters of the VC in Fig. 43 can be significantly improved if you include at its input a parallel buffer cascade that has proven itself well with “twos” (Fig. 44).

Variant of VK Siklai according to the diagram in Fig.

4 g using composite Bryston transistors is shown in Fig. 46. In Fig. Figure 48 shows a variant of the VK on Sziklai transistors (Fig. 4e) with a transmission coefficient of about 5, in which the input transistors operate in class A (thermostat circuits are not shown).

Anti-saturation circuits of output transistors contribute to increasing the reliability of amplifiers by eliminating through currents, which are especially dangerous when clipping high-frequency signals. Variants of such solutions are shown in Fig. 58. Through the upper diodes, excess base current is discharged into the collector of the transistor when approaching the saturation voltage. The saturation voltage of power transistors is usually in the range of 0.5...1.5 V, which approximately coincides with the voltage drop across the base-emitter junction. In the first option (Fig. 58 a), due to the additional diode in the base circuit, the emitter-collector voltage does not reach the saturation voltage by approximately 0.6 V (voltage drop across the diode). The second circuit (Fig. 58b) requires the selection of resistors R 1 and R 2. The lower diodes in the circuits are designed to quickly turn off the transistors during pulse signals. Similar solutions are used in power switches.

Often, to improve the quality, UMZCHs are equipped with separate power supply, increased by 10...15 V for the input stage and voltage amplifier and decreased for the output stage. In this case, in order to avoid failure of the output transistors and reduce the overload of the pre-output transistors, it is necessary to use protective diodes. Let's consider this option using the example of modification of the circuit in Fig. 39. If the input voltage increases above the supply voltage of the output transistors, additional diodes VD 1, VD 2 open (Fig. 59), and the excess base current of transistors VT 1, VT 2 is dumped onto the power buses of the final transistors.

In this case, the input voltage is not allowed to increase above the supply levels for the output stage of the VC and the collector current of transistors VT 1, VT 2 is reduced.

Bias circuits

Previously, for the purpose of simplicity, instead of a bias circuit in the UMZCH, a separate voltage source was used. Many of the considered circuits, in particular, output stages with a parallel follower at the input, do not require bias circuits, which is their additional advantage. Now let's look at typical displacement schemes, which are shown in Fig. 60, 61. Stable current generators. A number of standard circuits are widely used in modern UMZCHs: a differential cascade (DC), a current reflector ("current mirror"), a level shift circuit, a cascode (with serial and parallel power supply, the latter is also called a "broken cascode"), a stable generator current (GST), etc. Their correct use can significantly increase UMZCH. We will estimate the parameters of the main GTS circuits (Fig. 62 - 6 6) using modeling. We will assume that the GTS is a load of the UN and is connected in parallel with the VC. We study its properties using a technique similar to the study of VC.

Current reflectors

The considered GTS circuits are a variant of a dynamic load for a single-cycle UN. In an UMZCH with one differential cascade (DC), to organize a counter dynamic load in the UN, they use the structure of a “current mirror” or, as it is also called, a “current reflector” (OT). This structure of the UMZCH was characteristic of the amplifiers of Holton, Hafler, and others. The main circuits of the current reflectors are shown in Fig. 67. They can be either with a unity transmission coefficient (more precisely, close to 1), or with a greater or lesser unit (scale current reflectors). In a voltage amplifier, the OT current is in the range of 3...20 mA: Therefore, we will test all OTs at a current of, for example, about 10 mA according to the diagram in Fig. 68.

The test results are given in table. 3.

As an example of a real amplifier, the S. BOCK power amplifier circuit, published in the journal Radiomir, 201 1, No. 1, p. 5 - 7; No. 2, p. 5 - 7 Radiotechnika No. 11, 12/06

The author's goal was to build a power amplifier suitable for both sounding "space" during festive events and for discos. Of course, I wanted it to fit in a relatively small-sized case and be easily transported. Another requirement for it is the easy availability of components. In an effort to achieve Hi-Fi quality, I chose a complementary-symmetrical output stage circuit. The maximum output power of the amplifier was set at 300 W (into a 4 ohm load). With this power, the output voltage is approximately 35 V. Therefore, the UMZCH requires a bipolar supply voltage within 2x60 V. The amplifier circuit is shown in Fig. 1 . The UMZCH has an asymmetrical input. The input stage is formed by two differential amplifiers.

A. PETROV, Radiomir, 201 1, No. 4 - 12

Fig.3.1

This is the simplest design that allows you to demonstrate the amplification capabilities of a transistor. True, the voltage gain is small - it does not exceed 6, so the scope of such a device is limited. However, you can connect it to, say, a detector radio (it should be loaded with a 10 kΩ resistor) and use the BF1 headset to listen to broadcasts from a local radio station.

The amplified signal is supplied to input jacks X1, X2, and the supply voltage (as in all other designs of this author, it is 6 V - four galvanic elements with a voltage of 1.5 V each connected in series) is supplied to jacks X3, X4. Divider R1 R2 sets the bias voltage at the base of the transistor, and resistor R3 provides current feedback, which helps temperature stabilization of the amplifier.

How does stabilization occur? Let us assume that the collector current of the transistor increases under the influence of temperature. Accordingly, the voltage drop across resistor R3 will increase. As a result, the emitter current will decrease, and therefore the collector current will decrease - it will reach its original value.

The load of the amplifier stage is a headphone with a resistance of 60...100 Ohms.

It is not difficult to check the operation of the amplifier; you need to touch the input jack X1, for example, with tweezers - a faint buzzing sound should be heard in the phone, as a result of the alternating current. The collector current of the transistor is about 3 mA.

Fig.3.2

It is designed with direct coupling between stages and deep negative DC feedback, which makes its mode independent of ambient temperature. The basis for temperature stabilization is resistor R4, which “works” similarly to resistor R3 in the previous design.

The amplifier is more “sensitive” compared to a single-stage one - the voltage gain reaches 20. You can supply AC voltage amplitude of no more than 30 mV, otherwise distortion will occur that can be heard in the headphone.

They check the amplifier by touching the input jack X1 with tweezers (or just a finger) - a loud sound will be heard in the phone. The amplifier consumes a current of about 8 mA.

This design can be used to enhance weak signals, for example, from a microphone. And of course, it will significantly enhance the AF signal taken from the load of the detector receiver.

This book discusses the features of circuit solutions used in the creation of miniature transistor radio transmitting devices. The relevant chapters provide information on the principles of operation and features of the functioning of individual units and cascades, circuit diagrams, as well as other information necessary for the independent construction of simple radio transmitters and radio microphones. A separate chapter is devoted to the consideration of practical designs of transistor microtransmitters for short-range communication systems.

The book is intended for beginning radio amateurs interested in the features of circuit design solutions for units and cascades of miniature transistor radio transmitting devices.

In miniature transistor radio transmitting devices there is often a need to obtain of great importance low-frequency signal gain, which requires the use of two or more amplification stages. In this case, the use of multi-stage capacitively coupled microphone amplifiers, each of the stages of which is made on the basis of the considered circuits, does not always lead to satisfactory results. Therefore, circuit solutions for microphone amplifiers with direct coupling between cascades have become widespread in miniature radio transmitting devices.

Such amplifiers contain fewer parts, have lower energy consumption, are easy to configure and are less critical to changes in the supply voltage. In addition, amplifiers with direct coupling between stages have a more uniform bandwidth, and nonlinear distortion they can be minimized. One of the main advantages of such amplifiers is their relatively high temperature stability.

However, high temperature stability, like the other advantages of amplifiers with direct coupling between stages listed above, can only be realized by using deep negative DC feedback supplied from the output to the first stage of the amplifier. When using the appropriate circuit design, any current changes caused by both temperature fluctuations and other reasons are amplified by subsequent stages and fed to the amplifier input in this polarity. As a result, the amplifier returns to its original state.

Schematic diagram One of the options for a two-stage microphone amplifier with direct coupling between stages is shown in Fig. 2.11. With a supply voltage of 9 to 12 V and a maximum input voltage of 25 mV, the output voltage level in the frequency range from 10 Hz to 40 kHz can reach 5 V. In this case, the current consumption does not exceed 2 mA.

Rice. 2.11. Schematic diagram of a microphone amplifier with direct coupling between stages (option 1)

The low-frequency signal generated by microphone VM1 is fed through the isolation capacitor C2 to the input of the first amplifier stage, made on transistor VT1. Capacitor C1 filters unwanted high-frequency components of the input signal. Through resistor R1, supply voltage is supplied to the electret microphone VM1.

The amplified signal from the collector load of transistor VT1 (resistor R2) is supplied directly to the base of transistor VT2, on which the second amplifier stage is made. From the collector load of this transistor, the signal goes to the output of the amplifier through the isolation capacitor C4.

It should be noted that resistor R2, used as a load resistor in the collector circuit of transistor VT1, has a relatively high resistance. As a result, the voltage at the collector of transistor VT1 will be quite low, which allows you to connect the base of transistor VT2 directly to the collector of transistor VT1. The resistance value of resistor R6 also plays a significant role in choosing the operating mode of transistor VT2.

A resistor R4 is connected between the emitter of transistor VT2 and the base of transistor VT1, which ensures the occurrence of negative direct current feedback between the cascades. As a result, the voltage at the base of transistor VT1 is formed using resistor R4 from the voltage present at the emitter of transistor VT2, which in turn is formed when the collector current of this transistor passes through resistor R6. For alternating current, resistor R6 is shunted by capacitor C3.

If for some reason the current passing through transistor VT2 increases, then the voltage across resistors R5 and R6 will correspondingly increase. As a result, thanks to resistor R4, the voltage at the base of transistor VT1 will increase, which will lead to an increase in its collector current and a corresponding increase in the voltage drop across resistor R2, and this will cause a decrease in the voltage at the collector of transistor VT1, to which the base of transistor VT2 is directly connected. Reducing the voltage value at the base of transistor VT2 will lead to a decrease in the collector current of this transistor and a corresponding decrease in the voltage across resistors R5 and R6. At the same time, the voltage at the base of transistor VT1 will decrease, this transistor will shut down and will again operate in the normal, originally set mode. Thus, the currents and operating points of transistors VT1 and VT2 will be stabilized. The stabilization circuit operates in a similar way when the collector current of transistor VT2 may decrease, for example, when the ambient temperature decreases.

For amplifiers with direct coupling between stages, to set the mode it is usually enough to select the resistance value of only one resistor. In the considered circuit, the operating mode is set by selecting the resistance of resistor R6 or resistor R2.

Due to the fact that resistor R3 is not bypassed by a capacitor, AC feedback occurs in this amplifier, providing a sharp reduction in distortion.

It should be noted that with any change in the value of resistor R4 or the value of the amplifier supply voltage, it is necessary to adjust the position of the operating point. An important role in this process is played by resistor R6, instead of which, during the process of establishing the design, a trimming resistor is usually installed, which ensures the correct selection of the operating point of transistors VT1 and VT2.

A schematic diagram of another version of a two-stage microphone amplifier with direct coupling between stages is shown in Fig. 2.12. A distinctive feature of this circuit solution, compared to the previous one, is that to stabilize the operating mode, the proposed circuit uses two feedback circuits from output to input.

Rice. 2.12. Schematic diagram of a microphone amplifier with direct coupling between stages (option 2)

It is easy to see that in addition to transmitting the voltage removed from the emitter of transistor VT2 to the base of transistor VT1 through resistor R4, this design also ensures that the emitter voltage of the first stage transistor changes depending on the amount of current passing through the collector load of transistor VT2 (resistor R6). The second feedback circuit, connected between the collector of transistor VT2 and the emitter of transistor VT1, is formed by resistor R5 and capacitor C3 connected in parallel. It should be noted that the value of the upper limit frequency of the passband of a given microphone amplifier depends on the value of the capacitance of capacitor C3.

With a supply voltage of 9 to 15 V and a maximum input voltage of 25 mV, the output voltage level of the considered two-stage amplifier in the frequency range from 20 Hz to 20 kHz can reach 2.5 V. In this case, the current consumption does not exceed 2 mA.

A schematic diagram of another version of a microphone amplifier with direct coupling between stages is shown in Fig. 2.13.

Rice. 2.13. Schematic diagram of a microphone amplifier with direct coupling between stages (option 3)

In this design, the signal generated by microphone VM1 passes through the isolation capacitor C1 and resistor R2 to the base of transistor VT1, on which the first amplification stage is assembled. The amplified signal from the collector of transistor VT1 is supplied directly to the base of transistor VT2 of the second amplifier stage.

A resistor R4 is connected between the emitter of transistor VT2 and the base of transistor VT1, which ensures the occurrence of negative direct current feedback between the cascades. As a result, the voltage at the base of transistor VT1 is formed using resistor R4 from the voltage at the emitter of transistor VT2, which in turn is formed when the collector current of this transistor passes through resistor R6. For alternating current, resistor R6 is shunted by capacitor C3.

The signal generated at the collector of transistor VT2 is fed through the isolation capacitor C4 and potentiometer R8 to the output of the microphone amplifier. To reduce frequency distortion in the low-frequency region, the capacitance of the isolation capacitor C4 is increased to 20 μF. Potentiometer R8 performs the function of adjusting the level of the output low-frequency signal and has a logarithmic characteristic (type B).

In conventional amplifier stages, in which the transistor is connected in a circuit with a common emitter, the gain of the stage is determined primarily by the characteristics of the transistor itself. In this circuit, the gain largely depends on the parameters of the second feedback circuit connected between the amplifier output and the emitter of transistor VT1. In the circuit under consideration, this feedback circuit is formed by resistor R7. Theoretically, the gain K of a two-stage amplifier stage with direct coupling is determined by the ratio of the resistance values ​​of resistors R7 and R3, that is, it is calculated by the formula:

KUS = R7/R3.

For the cascade under consideration, the coefficient KUS = 10000/180 = 55.55. The above formula is valid for gain values ​​ranging from 10 to 100. For other ratios, additional factors come into force that affect the gain value. Special calculation methods should be used in cases where serial or parallel RC circuits are included in the feedback circuit.

Considering the classic circuits of microphone amplifiers based on bipolar transistors, one cannot fail to mention a two-stage amplifier made on two bipolar transistors of different conductivities. Schematic diagram of a simple microphone amplifier made on n-p-n and pnp transistors, shown in Fig. 2.14.

Rice. 2.14. Schematic diagram of a microphone amplifier using bipolar transistors of different conductivities

Despite its simplicity, this amplifier, which can be used to amplify signals taken from the output of a condenser microphone, has very acceptable parameters. With a supply voltage of 6 to 12 V and a maximum input voltage of 100 mV, the output voltage level in the frequency range from 70 Hz to 45 kHz reaches 2.5 V.

The signal generated at the output of the microphone VM1 is fed through the isolation capacitor C1 to the base of the transistor VT1, which has n-p-n conductivity, on which the first amplifier stage is made. The bias voltage supplied to the base of transistor VT1 is generated by a divider, which is formed by resistors R2 and R3.

Size of decline frequency response of a given microphone amplifier in the low-frequency region largely depends on the capacitance of the coupling capacitor C1. The smaller the capacitance of this capacitor, the greater the drop in frequency response. Therefore, with the capacitance value of capacitor C1 indicated in the diagram, the lower limit of the range of frequencies reproduced by the amplifier is at a frequency of about 70 Hz.

From the collector of transistor VT1, the amplified signal is supplied directly to the base of transistor VT2, which has p-n-p conductivity, on which the second amplifier stage is made. This amplifier, as in the previously discussed designs, uses a circuit with direct coupling between stages. Resistor R4, which has a high resistance, is used as a load resistor in the collector circuit of transistor VT1. As a result, the voltage at the collector of transistor VT1 will be relatively small, which allows the base of transistor VT2 to be connected directly to the collector of transistor VT1. The resistance value of resistor R7 also plays a significant role in choosing the operating mode of transistor VT2.

The signal generated at the collector of transistor VT2 is fed through the isolation capacitor C4 to the output of the microphone amplifier. To reduce frequency distortion in the low-frequency region, the capacitance of the isolation capacitor C4 is increased to 10 μF. The magnitude of the decline in the high-frequency region of the range reproduced by the amplifier can be achieved by reducing the load resistance, as well as by using transistors with a higher limiting frequency.

The gain of this amplifier is determined by the ratio of the resistances of resistors R5 and R6 in the feedback circuit. Capacitor C3 limits the gain by higher frequencies, preventing the amplifier from self-excitation.

When using a condenser microphone, the voltage required to power it will need to be supplied to its switching circuit. For this purpose, resistor R1 is installed in the circuit, which is also a load resistor for the microphone output. When using the microphone amplifier in question with an electrodynamic microphone, resistor R1 can be excluded from the circuit.

Particularly noteworthy are the circuit solutions of two-stage microphone amplifiers, in which the input stage is made of a field-effect transistor, and the output stage is made of a bipolar transistor. Schematic diagram of one of the options for a simple microphone amplifier, made on a field and bipolar transistors, shown in Fig. 2.15. This design is characterized not only by a low noise level and a relatively high input impedance, but also by a significant frequency range of the amplified signal. With a supply voltage of 9 to 12 V and a maximum input voltage of 25 mV, the output voltage level in the frequency range from 10 Hz to 100 kHz can reach 2.5 V. In this case, the current consumption does not exceed 1 mA, and the input resistance is 1 MOhm.

Rice. 2.15. Schematic diagram of a microphone amplifier using field-effect and bipolar transistors of different conductivity

The signal taken from the output of microphone VM1 is fed through the isolation capacitor C1 and resistor R1 to the gate field effect transistor VT1, on which the input amplifier stage is made. Resistor R2, the value of which determines the value of the input resistance of the entire structure, provides direct current connection between the gate of transistor VT1 and the housing bus. For direct current, the position of the operating point of transistor VT1 is determined by the resistance values ​​of resistors R3, R4 and R5. For alternating current, resistor R5 is shunted by capacitors C2 and C3. The relatively large capacitance of capacitor C2 provides sufficient gain in the lower part of the frequency range of the amplified signal. In turn, the capacitance value of capacitor C3 provides sufficient gain in the upper part of the frequency range.

The amplified signal is removed from load resistor R3 and supplied directly to the base of transistor VT2, which has p-n-p conductivity, on which the second amplification stage is made. Resistor R6, included in the collector circuit of transistor VT2, is not only a load resistor in the second amplifier stage, but is also part of the feedback circuit of transistor VT1. The ratio of the values ​​of resistors R6 and R4 determines the gain of the entire structure. If necessary, the gain can be reduced by selecting the resistance value of resistor R4. The signal generated at the collector of transistor VT2 is fed through resistor R7 and separating capacitor C4 to the output of the microphone amplifier.

The simpler the design, the more room for creativity there is. The two-stage amplifier circuit is polished to a shine, but you can “season” the sound to your own taste.

HOLY SIMPLICITY

This material, unlike most others, was not ordered by the editors, but arrived “by gravity” by e-mail. Therefore, there will be no traditional presentation of the author - with a portrait and compliments. We are sure that after reading it, a portrait will be drawn in your imagination, but decide for yourself about compliments.

Intro

Actually, sound is a matter of taste. From the circuit, I tried to achieve neutrality, detail, and aurally even timbre and frequency balance, as a starting point for further procedures. Sort of like a blank canvas.

By detail, I mean the transmission of subtle shades of timbre, reverberation, natural attenuation of sounds, after-sound... It, detail, is manifested in the naturalness of transmission and natural dynamics of sounds that are well known to us, absorbed by us since childhood.

As for music, here, especially on poorly made recordings, sometimes you want to touch up something or, conversely, cover it up. Up to setting the switch “soft - neutral - dynamic”.

As a result, all solutions were finally selected (or rejected) by listening. This is my amp and it sounds the way I think it should. Without claims to Absoluteness...

At the same time, I didn’t really focus on the fact that the scheme would not tolerate free intervention and would not be suitable for dummies with modest incomes. But, despite its apparent simplicity, the amplifier circuit took a long time to polish - several years. Its capabilities will be revealed only with a good source and acoustics.

To my ears, the amplifier came out of the soldering iron transparent enough to get any desired type of sound by selecting the appropriate parts. If one of you or your friends at least tries the first cascade (in fact, the whole highlight is in it!) in the most strict environment, it would be absolutely great! And then references to rave reviews from only one person, moreover, the author of the scheme, are not entirely convincing.

First of all, this is the anode resistor of the first stage and the interstage capacitor. Well, the other components also mean something...

Part 1

Here we go! The input signal is supplied to the grid of lamp L1 through the anti-ringing choke Dr1. The choice of a choke instead of a traditional resistor is explained, first of all, by its better sound properties in comparison with a conventional resistor. It should also be noted that the 6S17K lamp exhibits instability at HF. The choke eliminates self-generation without introducing noticeable distortion. Of course, using a regular 1 kOhm resistor also solves this problem, but slightly spoils the sound.

The first stage is built according to a fixed-bias circuit. The construction of the circuit was determined by the following “technical specifications”:

Refusal of the shunt capacitor in the cathode circuit;

Refusal of unwanted negative feedback in the same circuit through a “classical” resistor;

Rejection of the first transfer capacitor;

Operation from a music signal source with zero DC component at the output.

Thus, it was impossible to assign the task of organizing the grid displacement to the signal source. A circuit was developed and tested with a cathode resistor of a very small value (from fractions to a few Ohms), the required voltage drop across which was obtained not due to the lamp cathode current, as in the classical circuit, but by supplying this resistor with a large current from a separate source. In practice, such a source was a +6.5 V filament stabilizer.

Initially, the required current was set by an external resistor, the value of which was determined from the required bias voltage at the cathode. In a specific circuit, it turned out to be possible to use the filament current of the 6S17K-V lamp itself (300 mA), especially since one of the terminals of the filament is connected to the cathode inside the lamp. There were many doubts about the quality of the circuit's operation, there were concerns about interference from the filament stabilizer getting into the amplified signal, but everything turned out to be good.

The filament regulator is nothing special: a low-dropout diode bridge rectifier, a 10,000uF/16V electrolytic capacitor, and a 7806 silicon diode regulator in series with common to boost the voltage from 6 to 6.5V.

The sound turned out to be clearly better than in circuits with a grid input and/or bypass cathode capacitor, regardless of the quality of these capacitors. Over the course of a year, I returned twice to “classical” circuits with capacitors in the indicated places and was always convinced of their inferiority.

Undesirable feedback on the cathode resistor is also practically absent due to its small value.

I will not insist on the absolute novelty of this solution, but let anyone who finds another amplifier circuit with such a trick throw a stone at me!

Part 2

In principle, under “normal” conditions, any input lamps with a low bias voltage can be used. In this case, it is better to set the bias current with a separate resistor, and not to drive it through the filament, as I did. But this will not affect the sound - checked. I tried all sorts of tubes, ranging from 6S2P, 6S3P and ending with exotics such as 6S53N Nuvistors or subminiature triodes, but the gain was still sorely lacking. Along the way, I found out that the advertised 6S45P tube is actually not that good - the sound is muddy and blurry. But the 6S3(4)P is wonderful, and the nuvistors are simply magnificent! From the experience of friends and acquaintances, I can also say that for 2S4S with a traditional input, you can stop at 6Zh4 (foreign analogues - 6AS7, 6F10, 6AJ7) in triode connection and an interstage transformer.

It is possible with a larger bias, such as 6H8C, but the voltage of the auxiliary source will have to be raised to 30 volts, which is inconvenient.

My final choice of a lamp for the input stage was determined by several requirements. Firstly, I wanted to limit myself to a simple two-stage amplifier circuit. Secondly, to obtain a sensitivity of no worse than 0.15 - 0.2 V, since the input stage of the amplifier was supposed to work directly with the signal coming from the current output of the DAC.

The DAC is very simple: an AD1860 converter, the current output of which goes to a 619 Ohm resistor. It is this resistor that is designated in the diagram as R1. No filters. The DAK box (formerly DAC-in-BOX Audio Alchemy) is placed directly in the amplifier case, the wires from the box are routed to the input lamp, and resistor R1 is immediately connected. The idea was this: get the current as far away from the DAC as possible in order to be insensitive to the nonlinearities of the contacts and soldering, and solder the resistor I-U converter right next to the entrance lamp. By the way, the resistor is a leadless type C6-9 with dimensions of approximately 1 x 1 x 1.5 mm.

And then in the reference book a previously unknown lamp 6S17K-V was discovered. At first I flipped through it without looking, deciding that this was another generator product with the “right” characteristic. In addition, the connection of the filament and cathode inside the cylinder made it unsuitable for almost all standard connections, which apparently explains its complete absence in sound amplification circuits. The impossibility of installing this lamp in a socket apparently also scared amplifier builders away from it. And the last nail in the lid was hammered in with the ridiculous operating time figure of 200 hours, according to the reference book.

But then reason prevailed, and the following things became clear:

1. The lamp fits my bias design perfectly.
2. A gain of about 150 - 180 allows you to achieve the desired sensitivity with two stages.
3. The durability of the insert for this lamp is actually 2000 hours, and taking into account its underload in terms of power (1.2 W at a maximum of 2) and undervoltage filament (5.7 V, as can be easily calculated by looking at the diagram), one can expect that its service life will be no worse than that of electrolytic capacitors.
4. Direct mounting has a beneficial effect on the sound due to the absence of unnecessary contacts, wires and soldering.
5. In a real circuit, the lamp is very linear, and specifically in my circuit there is a margin of 6 - 8 dB for overload before audible distortion appears. Moreover, this can be judged when the volume control is turned on like mine, but this is getting a little ahead of ourselves.
6. There is a fly in the ointment: lamps have a wide range of parameters...
7. ...but another bucket of honey: the lamp does not suffer from the microphone effect, despite the high slope (10 mA/V) and a gain of about two hundred.

Yes, it won’t work with vinyl, and it won’t work with a good tape recorder either - there’s no input headroom. Even, in general, back to back, and for a DAC the gain is crazy. And there are also input trances... But, despite the seeming frailty of the 6S17K-V as a driver, everything is much better than one might expect. I have not noticed any instability in the 2S4S mode. The output impedance of the volume control is a maximum of 25 kOhm in the middle position, a fairly small value. And no one bothers to reduce the leakage resistor at least ten times with a corresponding increase in the interstage capacitance. In the end, we are talking about a specific and workable scheme.

So my attempt to create a “Swedish family” between 6S17K-V, DAC and 2S4S turned out to be quite successful! And now, while you are reading these lines, everything is working great. Moreover, without audible distortion, despite the full swing at the input. I listen to it every evening. Probably, reference data and reality, as in Odessa, are two big differences.

However, I will repeat once again that if such amplification is not required, it is quite possible to install something more traditional, almost without changing the circuit. If one of you decides to use it, he will, of course, make changes to it in accordance with his vision and requests. In this case, it is better to move the volume control to its usual place - to the input. And that's all - it will work with any source!

Part 3

The amplified signal is removed from the anode load resistor R2, lamp L1 and goes to the volume control, made on a variable resistor R4.

Previously, I considered three options for turning on the volume control:

1. In parallel with the anode resistor R2. The disadvantages are obvious: when adjusting, there is a short-term change in the amplifier's DC mode, and almost certainly in sound signal There will be rustling noises from the engine. In addition, I was alarmed by the opinion of Seryozha Rubtsov about the inadmissibility of applying any significant constant bias to this type of resistor.
2. The resistor is grounded through a decoupling capacitor. This is what was done in my scheme. Black Gate (C2) shunted with fluoroplastic (C3) is used as a decoupler. There is a slight decrease in the maximum voltage swing, which can be easily compensated by increasing the supply voltage. That is why it is higher in the first cascade than in the second.
3. The resistor is directly grounded. The disadvantages are similar to point 1. Moreover, due to the formation of a divider R2/R4, the maximum voltage swing of the first stage is sharply reduced. It won't work, although the absence of a capacitor could theoretically improve the sound.

Firm "ERAudio" (formerly "NEM"), Novosibirsk. - Approx. ed.

Moving the regulator from the input circuits to the middle of the circuit is explained simply: its negative impact on the sound is too strong, despite the high cost and attempts to turn it on using the G-regulator circuit. The uncompromising construction of the first stage seemed to push the volume control into the high-current sections of the circuit. I’ll say right away that such a construction is possible only if there is a guarantee of the absence of voltage overloads of the first stage. This is not a problem with a digital source (you can’t jump above 0 dB), but, for example, with a tape recorder you will need to be careful. With vinyl or an arbitrary source, you will have to return the regulator to standard seat to the beginning of the circuit or provide for such sources an adjustable (or unregulated) attenuator at the corresponding input.

If bypassing is not required for the anode supply capacitor of the first stage C1, then for C2 it is desirable. I explain it this way: the low internal resistance of lamp L1 (several kOhm) with the high resistance of the anode load R2 form a divider that effectively cuts off possible dirty tricks from capacitor C1 from the amplified signal. That is, the signal is mainly determined by the lamp.

If the regulator is positioned at the beginning of the sector, the influence of C2 may be significant. Practice has shown that this is so. Even Black Gate isn't perfect! The influence is manifested primarily in a weak but noticeable sharpness of the top, as well as in some of their collapse. As it warms up (not “esoteric”, but the temperature itself) for about an hour, these effects weaken significantly, and the sound improves and noticeably “naturalizes”.

Perhaps the Black Gate "K/FK" series of capacitors, which are specifically designed for use in audio circuits and feature low noise levels of less than 150 dB, should be used. - Approx. ed.

The “esoteric” heating of the capacitor is associated, first of all, with the molding process, which occurs each time to one degree or another after voltage is applied to the electrodes. - Approx. ed.

Why this is so can be found in Klaus (www.klausmobile.narod.ru). He has a link to studies of nonlinearities and losses of capacitors, where it is very clearly shown how much (how many times!) the characteristics of electrolytic capacitors improve when heated.

The choice of the type of shunt capacitor is a question that I have not yet fully resolved, but it is not a big one: either fluoroplastic or paper-oil. Maybe also mica. That's all. No other films “roll” - I already understood that. The issue with the “oil” has not been resolved due to the lack of the necessary capacitors. The experiments are not finished, the process is ongoing...

Part 4

From the volume control, through the coupling capacitor C4, the signal is supplied to the 2C4C grid. There is no anti-ringing resistor, since my experiments showed its complete uselessness. The construction of the second cascade has no special features, except that instead of a powerful variable resistor to organize an artificial midpoint in the cathode to minimize the background, two constant resistors are used. Experience has shown that it is quite sufficient to use fixed resistors with a tolerance of no worse than 1%. The high quality of this solution is obvious, and there are no problems with the background, at least with 2S4S.

The type of resistors is not very critical here. They can be wire, metal film precision types. We only need to avoid carbon and all MLTs. A small nominal value with a low gain and slope of 2C4C do not create significant OOS on these resistors, which, in turn, does not require the use of special measures to remove this OOS.

You can notice that the lamps in my circuit are used with some power overload at the anode. This is out of greed, don’t pay attention, especially since nothing has been done to the lamps in over a year.

Resistors R8, R9 and R10 are designed to cut off possible nonlinearities in the output capacitors of the power supply from the amplifier. Again, this is explained by the formation of a divider consisting of the internal resistance of the Black Gate in the amplifier (no more than tens of mOhms) and the above-mentioned resistors themselves. In addition, these resistors significantly reduce inductive interference that can occur when external closed loops of connecting wires are formed. I have not yet conducted special experiments to identify the influence of these resistors on the sound.

At the end of the low-fatigue path, the signal from the 2C4C anode reaches the primary of the output transformer, the features of which can only be noted high quality and a very “bad” price. I evaluate its quality very simply: it is completely “transparent” to sound, its presence in the path is imperceptible. Any, even the most minor changes in the circuit, including unnecessary soldering and even the movement of the mounting wire, immediately become audible in my speakers.

If you look closely at the diagram, you will notice that the common filament wire of the first stage and the common terminal of capacitors C6 + C7 are not connected directly to the common point. This is not a coincidence, but I won’t say anything about the reasons for now. There must be some secrets left...

About tasty and healthy food

I made the power supply remote with separate power supply for the filament, preliminary circuits and final stage. It connects to the amplifier via a huge military grade connector with silver plated contacts. All main voltages of the unit, except for filament ones, are regulated, for which the simplest stabilizers on high-voltage field-effect transistors are used. Don’t swear that the power supply is diode! But with measures taken to suppress noise in general and reduce interference from diodes in particular.

“...If you have separate power supplies for the first and second stages, then you can quite easily do without an isolation capacitor. You connect the output lamp grid directly to the input anode (there is a constant potential of +200 volts), and from a low-current power source - from which the first stage is powered - using a high-resistance resistive divider you obtain a potential of +245 volts, and to this point you connect the cathode of the first lamp. Powerful power supply, fortunately it is isolated, is connected with the minus to the cathode of the output lamp, and with the plus to the “cold” end of the transformer. As a result, you get rid of the transfer capacitor and the entire fixed bias circuit. Two resistors and a high-voltage (unfortunately) capacitor are added, which bypasses the “ground” arm of the resistive divider. The method you used to control the volume is also appropriate in this configuration.” - Approx. sympathetic Andrei from the Internet.

At the output of the power supply there are “soft” stabilizers according to the simplest scheme: field worker in repeater mode, in the gate circuit there is a semiconductor zener diode. Zener diode output via series resistor hooked up to a large capacitance, connected at the other end to a common wire - it gives a smooth start and finishes off possible pulsations, interference, and noise. There is a variable resistor in parallel with the capacitor, the motor of which is connected to the gate. All!

High-voltage pulse rectifier diodes. Any suitable for switching power supplies with a permissible reverse voltage of at least three times the rectified voltage will be suitable here. Now all this is easy to find on any radio market. Specifically, the K20-39 were simply at hand.

There are 10 Ohm resistors in series with the diodes, and a 0.1 µF ceramic capacitance in parallel to the diodes (parallel to each). At the input of the rectifier there is a capacitance of 0.1 mF, at the output - 1.0 μF.

The filament transformer is TPP 304, the low-power anode transformer (for powering the preliminary stage) is TA 84-220-50, the high-power anode transformer is TS180. Anode transformers are connected to the network through a noise suppression filter. As a result, the amplifier turned out to be completely insensitive to interference from the network, even to the clicks of an old refrigerator.

Picked out from the monitor, it is a C-L-C filter.

I have plans to order or purchase branded foreign transformers, otherwise domestic products do not inspire confidence - they hum.

You can also try custom “Electron-Complex” ones. - Approx. ed.

Of course, I made a mock-up of the power supply using the 5Ts3S and 6Ts4P kenotrons. Well, where would I be without this! As seditious as it may be, in my circuit it did not show any noticeable advantages over a semiconductor power supply. Perhaps the fact is that both power supplies used large output capacitances of 470 μF, and I was able to effectively get rid of the interference of the diode bridge. In addition, the stabilizer, being simply a source follower, is completely indifferent to load variability. So I had to put the kenotron power supply away and forget about it, since the voltage in my outlet fluctuates freely from 170 to 220 V. In any case, thanks to our military industry, changing blocks takes a minute.

Part 5

In terms of sound, the circuit turned out to be very sensitive to the quality of installation and the amount of soldering, so much so that the cathode circuit had to be radically minimized: the cathode resistor was soldered with one terminal directly to the lamp leg, and the other to the common point of the circuit. The installation of the input stage and volume control circuits is made with Jensen silver monocore with a diameter of 0.8 mm. All other circuits are made of copper wire.

Also this scheme very sensitive to the type of cathode resistor. Carbon ones, including BLP, turned out to be simply disgusting, wire ones were satisfactory, but nothing more. I didn’t really like the PTMN at all, although I collected a monstrous number of them for experiments. As a tuning element for obtaining the desired color of the sound of the amplifier as a whole, the cathode resistor is unsuitable.

The anode resistor of the first stage is the ideal element for the necessary touch-up of the amplifier's sound! The choice of the type of this resistor has a direct impact on the sound.

Currently I have this tantalum foil resistor, but I still haven’t been able to make a final choice between it and the Riken Ohm. Their sound is different: Riken Ohm gives a very beautiful color to the mids, some special dynamics, softening the top and slightly blurring the detail, while tantalum is sterile and very detailed.

It was with tantalum resistors that I was ambushed. About a year ago, having poured out my thoughts on the Internet (www.dvdworld.ru/cgi-bin/audiobbs.pl) about the sound qualities of various resistors, I rejected tantalum. But my later research showed that this was a trap, which I myself warned against falling into. The fact is that a good component may seem “bad” if, as a result of its installation in the circuit, the shortcomings of other path components appear. And the sharpness of the sound, which then seemed to me to be a property of tantalum, actually turned out to be a drawback of my then DAC. Now justice has prevailed, but I still like the sound of Riken Ohm.

In the leakage of the first stage, it is better to use something film - good and precise. It’s not by chance that I’m talking about precision resistors. Usually this means increased quality of the resistor in general. (In the second stage it is not so critical - you can use both film and carbon.) I suspect that tantalum or copper foil will be even better, but so far I have not been able to find them for such low denominations. The best here so far have been domestic C2-10.

S2-10 are high-frequency accurate, which is clearly visible upon external inspection. Main features:

• Shiny, unpainted caps.
• There are no spiral grooves on the conductive layer - non-inductive.
• There are traces of adjustment - longitudinal cuts made with a diamond blade.
• Some resistors have a dark bluish metallic tint to the conductive layer coating.

As for the choice of capacitor C4, my choice of FT is simply determined - this is the best of what I have tried. I can say the same for FT as for tantalum resistors: neutrality and detail without poison or harshness. I won't say that they are the best at all. For example, I really want to try the famous Jensen copper capacitors (paper - oil), which S. Rubtsov and O. Khavin spoke very positively about. As we say: “If there is money, there will be copper and oil!”

The following capacitors were auditioned: MBM, K40-U9, K73, K71 - everything is very bad! MultiCap RTX and PPFX, aluminum Jensen (paper - oil) 1973, SSG, K31 - passable, but no more.

The failure with the Jensen was probably due to the fact that they were old and purely electrical, despite being ripped out of some Audio Note.

If you are planning to build an amplifier, then I strongly recommend planning the costs of output transformers as follows:

1. Having a certain amount of money to build an amplifier and intending to spend it more or less immediately, set aside half and no less for transformers.
2. If you plan to spend a certain amount over a long period of time (gradual development), then increase the cost of trannys to two-thirds of this amount. Spending becomes easier over time.

Output transformers (or any transformers in general!) are never too good, there is simply not enough money. Even if you install cheap hardware in a good and “correct” circuit, a miracle will not happen, it will not play as well as it could. The transformer is the heart of the amplifier.

Unfortunately, serious technology for manufacturing high-quality transformers, especially for single ended amplifiers, over the last 80 years, has not come up with cheap solutions. So I don’t advise you to flatter yourself with the hope of winding a high-quality output transformer yourself in the kitchen. By the time you become more or less bearable with them, age-related diseases will have already set in, including hearing loss.

The production of truly good transformers is within the power of well-coordinated teams, for example, our native ERAudio from Novosibirsk or foreign guys from Tamura-Magnequest-Sowter, etc. At the same time, I would like to remind you once again the story that Tango transformers were no longer produced according to due to the advanced age of the Japanese grandfathers who made them, who were never able to pass on their accumulated experience to the younger generation.

Currently, Tango transformers continue to be produced in Japan, but by a different “team of authors.” Their range has thinned out by more than two-thirds, and expensive and high-quality single-cycle models have completely disappeared from it. Tango transformers of previous years are now gradually becoming antiques, including in terms of price. - Approx. ed.

Final

If the editors deem it possible, then a continuation will follow! In this case, I plan to tell the background and give several circuit options tested, a fixed-bias output stage circuit. I will also think about the optimal configuration of the amplifier, based on different budgets.

I've already figured it out. - Approx. ed.

Table 1